Space-time coded OFDM system for MMDS applications

ABSTRACT

A communications system includes a base station that is configured to transmit a signal that is modulated according to a predetermined modulation scheme and an orthogonal frequency division multiplexing scheme, wherein the signal is encoded using space-frequency coding. The base station includes a plurality of multiple-input multiple-output (MIMO) transceivers. The system includes a terminal that is configured to receive the modulated signal. The above arrangement is particularly applicable to providing multichannel multipoint distribution services (MMDS) over a radio communications system.

CROSS-REFERENCES TO RELATED APPLICATION

This application is related to, and claims the benefit of the earlierfiling date of U.S. Provisional Patent Application 60/246,023, filedNov. 6, 2000, entitled “Space-Time Codes OFDM System for MMDSApplications,” the entirety of which is incorporated herein byreference.

BACKGROUND OF THE INVENTION

1. Field of the Invention

The present invention relates to coding in a communications system, andis more particularly related to space-time codes that exploit multipleforms of diversity.

2. Discussion of the Background

Given the multiple-input multiple-output (MIMO) transceivers constantdemand for higher system capacity of wireless systems, multiple antennasystems have emerged to increase system bandwidth vis-a-vis singleantenna systems. In multiple antenna systems, data is parsed intomultiple streams, which are simultaneously transmitted over acorresponding quantity of transmit antennas. At the receiving end,multiple receive antennas are used to reconstruct the original datastream. To combat the detrimental effects of the communication channel,communication engineers are tasked to develop channel codes thatoptimize system reliability and throughput in a multiple antenna system.

To minimize the effects of the communication channel, which typically isRayleigh, space-time codes have been garnered significant attention.Rayleigh fading channels introduce noise and attenuation to such anextent that a receiver may not reliably reproduce the transmitted signalwithout some form of diversity; diversity provides a replica of thetransmitted signal. Space-time codes are two dimensional channel codesthat exploit spatial transmit diversity, whereby the receiver canreliably detect the transmitted signal. Conventional designs ofspace-time codes have focused on maximizing spatial diversity inquasi-static fading channels and fast fading channels. However, realcommunications systems exhibit channel characteristics that aresomewhere between quasi-static and fast fading. Accordingly, suchconventional space-time codes are not optimized.

Further, other approaches to space-time code design assume that channelstate information (CSI) are available at both the transmitter andreceiver. Thus, a drawback of such approaches is that the designrequires the transmitter and receiver to have knowledge of the CSI,which increases implementation costs because of the need for additionalhardware. Moreover, these approaches view the transmit diversityattending the use of space-time codes as a substitute for timediversity; consequently, such space-time codes are not designed to takeadvantage of other forms of diversity.

Notably, information theoretic studies have shown that spatial diversityprovided by multiple transmit and/or receive antennas allows for asignificant increase in the capacity of wireless communications systemsoperated in a flat Rayleigh fading environment [1] [2]. Following thisobservation, various approaches for exploiting this spatial diversityhave been proposed. In one approach, channel coding is performed acrossthe spatial dimension as well as time to benefit from the spatialdiversity provided by using multiple transmit antennas [3]. Tarokh etal. coined the term “space-time coding” for this scheme. One potentialdrawback of this scheme is that the complexity of the maximum likelihood(ML) decoder is exponential in the number of transmit antennas. Anotherapproach, as proposed by Foshini [5], relies upon arranging thetransmitted data stream into multiple independent layers and sub-optimalsignal processing techniques at the receiver to achieve performance thatis asymptotically close to the outage capacity with reasonablecomplexity. In this approach, no effort is made to optimize the channelcoding scheme.

Conventional approaches to space-time coding design have focusedprimarily on the flat fading channel model. With respect to thetreatment of MIMO frequency selective channels, one approach contendsthe that space-time codes that are designed to achieve a certaindiversity order in flat fading channels achieve at least the samediversity order in frequency selective fading channels. Such an approachfails to exploit the spatial and frequency diversity available in thechannel.

Based on the foregoing, there is a clear need for improved approachesfor providing a system that utilizes space-time codes that can beutilized in a MIMO selective fading channel. There is also a need todesign space-time codes that can exploit spatial diversity as well astime diversity. There is also a need to improve system reliabilitywithout reducing transmission rate. There is a further need to simplifythe receiver design. Therefore, an approach for employing space-timecodes that can enhance system reliability and throughput in a multipleantenna system is highly desirable.

SUMMARY OF THE INVENTION

The present invention addresses the above stated needs by providing amultichannel multipoint distribution service (MMDS) system that employsspace-time codes to transmit and receives signals under Line-of-Sight(LOS) and non-LOS operating conditions via multiple-inputmultiple-output transceivers. The system utilizes orthogonal frequencydivision multiplexing (OFDM) with a quadrature phase shift keying (QPSK)sub-carrier modulation, according to one embodiment of the presentinvention. The space-time coding, which may be adaptive, effectivelyexploits the use of the MIMO transceivers. Further, the MMDS systemprovides an automatic repeat requests (ARQ) error control mechanism.

According to one aspect of the present invention, a method is providedfor communicating over a radio communications system. The methodincludes encoding a signal using space-frequency coding, and modulatingthe encoded signal according to a predetermined modulation scheme and anorthogonal frequency division multiplexing scheme. The method alsoincludes transmitting the modulated signal to a terminal using aplurality of MIMO transceivers.

According to another aspect of the present invention, a device isprovided for communicating over a radio communications system. Thedevice includes an encoder that is configured to encode a signal usingspace-frequency coding. The device also includes a modulator that isconfigured to modulate the encoded signal according to a predeterminedmodulation scheme and an orthogonal frequency division multiplexingscheme. Further, the device includes a plurality of MIMO transceiversthat are coupled to the modulator. Each of the transceivers isconfigured to transmit the modulated signal to a terminal.

According to one aspect of the present invention, a communicationssystem for providing multichannel multipoint distribution services isdisclosed. The system includes a base station that is configured totransmit a signal that is modulated according to a predeterminedmodulation scheme and an orthogonal frequency division multiplexingscheme, wherein the signal is encoded using space-frequency coding. Thebase station includes a plurality of MIMO transceivers. The systemincludes a terminal that is configured to receive the modulated signal.

According to another aspect of the present invention, an apparatus isprovided for communicating over a radio communications system. Theapparatus includes means for encoding a signal using space-frequencycoding; means for modulating the encoded signal according to apredetermined modulation scheme and an orthogonal frequency divisionmultiplexing scheme; and means for transmitting the modulated signal toa terminal using a plurality of MIMO transceivers.

In yet another aspect of the present invention, a computer-readablemedium carrying one or more sequences of one or more instructions forcommunicating over a radio communications system is disclosed. The oneor more sequences of one or more instructions including instructionswhich, when executed by one or more processors, cause the one or moreprocessors to perform the step of encoding a signal usingspace-frequency coding. The encoded signal is modulated according to apredetermined modulation scheme and an orthogonal frequency divisionmultiplexing scheme, wherein the modulated signal is transmitted aterminal using a plurality of MIMO transceivers.

Still other aspects, features, and advantages of the present inventionare readily apparent from the following detailed description, simply byillustrating a number of particular embodiments and implementations,including the best mode contemplated for carrying out the presentinvention. The present invention is also capable of other and differentembodiments, and its several details can be modified in various obviousrespects, all without departing from the spirit and scope of the presentinvention. Accordingly, the drawing and description are to be regardedas illustrative in nature, and not as restrictive.

BRIEF DESCRIPTION OF THE DRAWINGS

A more complete appreciation of the invention and many of the attendantadvantages thereof will be readily obtained as the same becomes betterunderstood by reference to the following detailed description whenconsidered in connection with the accompanying drawings, wherein:

FIG. 1 is a diagram of a communications system configured to utilizespace-time codes, according to an embodiment of the present invention;

FIG. 2 is a diagram of an encoder that generates space-time codes, inaccordance with an embodiment of the present invention;

FIGS. 3A and 3B are diagrams of receivers that employ space-time codesand space-frequency codes, respectively, according to variousembodiments of the present invention;

FIG. 4 is a diagram of a communications system that supportsmultichannel multipoint distribution services using multi-inputmulti-output transceivers, according to the embodiments of the presentinvention;

FIG. 5 is a graph of the performance of a four transmit/four receiveantenna system employed in the system of FIG. 4, according toembodiments of the present invention;

FIG. 6 is a graph of the performance of a two transmit/one receiveantenna system, employed in the system of FIG. 4, according toembodiments of the present invention; and

FIG. 7 is a diagram of a computer system that can perform the processesof encoding and decoding of space-time codes, in accordance withembodiments of the present invention.

DESCRIPTION OF THE PREFERRED EMBODIMENTS

In the following description, for the purpose of explanation, specificdetails are set forth in order to provide a thorough understanding ofthe invention. However, it will be apparent that the invention may bepracticed without these specific details. In some instances, well-knownstructures and devices are depicted in block diagram form in order toavoid unnecessarily obscuring the invention.

FIG. 1 shows a diagram of a communications system configured to utilizespace-time codes, according to an embodiment of the present invention. Adigital communications system 100 includes a transmitter 101 thatgenerates signal waveforms across a communication channel 103 to areceiver 105. In the discrete communications system 100, transmitter 101has a message source that produces a discrete set of possible messages;each of the possible messages have a corresponding signal waveform.These signal waveforms are attenuated, or otherwise altered, bycommunications channel 103. One phenomena of interest is IntersymbolInterference (ISI), in which the channel 103 causes the overlap ofsignal pulses, resulting in the lost of signal orthogonality. Asdescribed with respect to the construction of space-frequency codes, thechannel ISI characteristics are minimized. It is evident that receiver105 must be able to compensate for the attenuation that is introduced bychannel 103.

To assist with this task, transmitter 101 employs coding to introduceredundancies that safeguard against incorrect detection of the receivedsignal waveforms by the receiver 105. To minimize the impact of thecommunication channel 103 on the transmission signals, channel coding isutilized. An algebraic design framework for layered and non-layeredspace-time codes in flat fading channels are in the following: A. R.Hammons Jr. and H. El Gamal. “On the theory of space-time codes for PSKmodulation,” IEEE Trans. Info. Theory, March 2000; and H. El Gamal andA. R. Hammons Jr. “The layered space-time architecture: a newprospective,” IEEE Trans. Info. Theory, 1999; each of which isincorporated herein by reference in its entirety.

Based upon the algebraic design framework for space-time coding in flatfading channels in “On the Theory of Space-Time Codes for PSKModulation,” A. R. Hammons Jr. and H. El Gamal, IEEE Trans. Info.Theory, March 2000, the present invention extends this framework todesign algebraic codes for multi-input multi-output (MIMO) frequencyselective fading channels. The codes, according to the presentinvention, optimally exploit both the spatial and frequency diversityavailable in the channel. Two design approaches with differentcomplexity-versus-diversity advantage trade-offs are considered. Thefirst approach (referred to as “single carrier time domain design”approach or STC (space-time coding)), which is more fully describedbelow in FIG. 3A, uses space-time coding and maximum likelihood (ML)decoding to exploit the multipath nature of the channel. The secondapproach utilizes an orthogonal frequency division multiplexing (OFDM)technique to transform the multi-path channel into a block fadingchannel (referred to as “OFDM based design” approach or SFC(space-frequency coding)); this approach is detailed in the discussionof FIG. 3B. The new algebraic framework, according to one embodiment ofthe present invention, is then used to construct space-frequency codesthat optimally exploit the diversity available in the resulting blockfading channel.

The two approaches, according to the present invention, differ in termsof decoder complexity, maximum achievable diversity advantage, andsimulated frame error rate performance. The first approach requiresrelatively greater complexity at the receiver 105 over the secondapproach, in that the first approach combines algebraic space-timecoding with maximum likelihood decoding to achieve the maximum possiblediversity advantage in MIMO frequency selective channels to achieve thediversity advantage. As a result, this first approach has a relativelylarge trellis complexity, as required by the maximum likelihood receiver105. The second approach utilizes an orthogonal frequency divisionmultiplexing (OFDM) front-end to transform an intersymbol-interference(ISD) fading channel into a flat block fading channel.

FIG. 2 shows a diagram of an encoder that generates space-time codes, inaccordance with an embodiment of the present invention. A transmitter200 is equipped with a channel encoder 203 that accepts input from aninformation source 201 and outputs coded stream of higher redundancysuitable for error correction processing at the receiver 105 (FIG. 1).The information source 201 generates k signals from a discrete alphabet,X′. Encoder 203 generates signals from alphabet Y to a modulator 205.Modulator 205 maps the encoded messages from encoder 203 to signalwaveforms that are transmitted to L_(t) number of antennas 207, whichemit these waveforms over the communication channel 103. Accordingly,the encoded messages are modulated and distributed among the L_(t)antennas 207. The transmissions from each of the L_(t) transmit antennas207 are simultaneous and synchronous.

FIG. 3A shows a diagram of a decoder that decodes space-time codes,according to an embodiment of the present invention. At the receivingside, a receiver 300 includes a demodulator 301 that performsdemodulation of received signals from transmitter 200. These signals arereceived at multiple antennas 303. The signal received at each antenna303 is therefore a superposition of the L_(t) transmitted signalscorrupted by additive white Gaussian noise (AWGN) and the multiplicativeintersymbol interference (ISI) fading. After demodulation, the receivedsignals are forwarded to a decoder 305, which attempts to reconstructthe original source messages by generating messages, X′ Receiver 300,according to one embodiment of the present invention, has a memory 307that stores channel state information (CSI) associated with thecommunication channel 103. Conventional communications systems typicallyrequire that CSI be available at both the transmitter and the receiver.By contrast, the present invention, according to one embodiment, doesnot require CSI at the transmitter 200, thus, providing a more robustdesign.

At the receiver 300, the signal r_(i) ^(j) received by antenna j at timet is given by

$r_{t}^{j} = {{\sqrt{E_{s}}{\sum\limits_{l = 0}^{L_{I\; S\; I} - 1}{\sum\limits_{i = 1}^{L_{t}}{\alpha_{l}^{i\; j}s_{t - 1}^{i}}}}} + n_{t}^{j}}$

where √{square root over (E)}_(s), is the energy per transmitted symbol;α_(t) ^(ij) is the complex path gain from transmit antenna i to receiveantenna j for the lth path; L_(ISI) is the length of the channel impulseresponse; s_(t) ^(i) is the symbol transmitted from antenna i at time t;n_(t) ^(j) is the additive white Gaussian noise sample for receiveantenna j at time t. The noise samples are independent samples ofcircularly symmetric zero-mean complex Gaussian random variable withvariance N₀/2 per dimension. The different path gains α_(t) ^(ij) areassumed to be statistically independent.

A space-time code is defined to include an underlying error control codetogether with a spatial parsing formatter. Specifically, an L_(t)×1space-time code C of size M has an (L_(t)l, M) error control code C anda spatial parser σ that maps each code word vector cεC to an L_(t)×lmatrix c whose entries are a rearrangement of those of c. The space-timecode C is said to be linear if both C and σ are linear.

It is assumed that the standard parser maps

$\overset{\_}{c} = {\left( {c_{1}^{(1)},c_{1}^{(2)},\ldots\mspace{11mu},c_{1}^{(L_{t})},c_{2}^{(1)},c_{2}^{(2)},\ldots\mspace{11mu},c_{2}^{(L_{t})},\ldots\mspace{11mu},c_{l}^{(1)},c_{l}^{(2)},\ldots\mspace{11mu},c_{l}^{(L_{t})}} \right) \in C}$to  the  matrix $c = \left\lbrack \begin{matrix}c_{1}^{1} & c_{2}^{1} & \cdots & c_{n}^{1} \\c_{1}^{2} & c_{2}^{2} & \cdots & c_{n}^{2} \\\vdots & \vdots & ⋰ & \vdots \\c_{1}^{L_{t}} & c_{2}^{L_{t}} & \cdots & c_{n}^{L_{t}}\end{matrix} \right\rbrack$

The baseband code word f(c) is obtained by applying the modulationoperator f on the components of c. This modulation operator maps theentries of c into constellation points from the discrete complex-valuedsignaling constellation Ω for transmission across the channel. In thisnotation, it is understood that c_(t) ^((i)) is the code symbol assignedto transmit antenna i at time t and

s_(t)^((i)) = f(c_(t)^((i))).

The diversity advantage of a space-time code is defined as the minimumabsolute value of the slope of any pairwise probability of error versussignal-to-noise ratio curve on a log-log scale. To maximize the spatialdiversity advantage provided by the multiple transmit antenna inquasi-static flat fading MIMO channels, the following rank criterion isutilized [3] [4]: for the baseband rank criterion, d=rank(f(c)−f(e)) ismaximized over all pairs of distinct code words c, eεC. Therefore fullspatial transmit diversity is achieved if and only if rank(f(c)−f(e))=L,for all pairs of distinct code words c, eεC. It should be noted that inthe presence of L_(r) receive antennas 303, the total diversityadvantage achieved by this code is L_(t)L_(r).

Space-time code constructions for frequency selective fading channels isbased on the concept that in an ISI (intersymbol interference)environment with L_(ISI) paths, a space-time system with L_(t) transmitantennas 207 is equivalent to a space-time system operating in flatfading channel with L_(t)L_(ISI) transmit antenna 207. However, in thisequivalent model the code word matrices are restricted to have a certainspecial structure. This structure is captured in the followingdefinition for the baseband code word matrix in ISI environments:

${f(c)}_{I\; S\; I} = \left\lbrack \begin{matrix}{f(c)} & \underset{\_}{0} & \cdots & 0 \\0 & {f(c)} & \cdots & 0 \\\vdots & \vdots & ⋰ & \vdots \\0 & 0 & \cdots & {f(c)}\end{matrix} \right\rbrack$

-   -   where c is the code word matrix as defined in (2) below, and 0        is the L_(t)×1 all zero vector. From the equivalent model, it is        clear that in the frequency selective fading channels,        space-time codes can be constructed to achieve L_(t)L_(ISI)        transmit diversity order. Therefore, the following baseband        design criterion for space-time codes in the ISI channel is        established: for ISI baseband rank criterion,        d=rank(f_(ISI)(c)−f_(ISI)(e)) is maximized over all pairs of        distinct code words c, eεC. Full transmit diversity in this        scenario is equal to L_(t)L_(ISI), and is achieved if and only        if rank(f_(ISI)(c)−f_(ISI)(e))=L_(t)L_(ISI) for all pairs of        distinct code words c, eεC.

Next, the binary rank criteria is developed; this criteria facilitatethe construction of algebraic space-time codes for BPSK (BinaryPhase-Shift Keying) and QPSK (Quadrature Phase-Shift Keying) modulatedsystems with an arbitrary number of transmit antennas 207 and channelimpulse response lengths. A new code word matrix c_(ISI) that capturesthe nature of the ISI channel is defined as follows:

$c_{I\; S\; I} = \left\lbrack \begin{matrix}c & \underset{\_}{0} & \cdots & 0 \\0 & c & \cdots & 0 \\\vdots & \vdots & ⋰ & \vdots \\0 & 0 & \cdots & c\end{matrix} \right\rbrack$

It is first observed that in generalf(c _(ISI))≠f(c)_(ISI),  (2)sincef(0)≠0

However, it is noted the diversity advantage only depends on differencesbetween code words rather than the code words themselves, and thusf(c _(ISI))−f(e _(ISI))=f(c)_(ISI) −f(e)_(ISI)

for any signaling constellation. The previous result is the key to thealgebraic space-time constructions developed in this section.

Attention is now turned to the development of BPSK modulated codes,which may be utilized in the communications system 100 of FIG. 1. ForBPSK modulation, elements in c are drawn from the field F={0,1} ofintegers modulo 2. The modulation operator/maps the symbol c_(t)^((i))εF to the constellation point

s_(t)^((i)) = f(c_(t)^((i))) ∈ {−1, 1}according to the rule

f(c_(t)^((i))) = (−1)^(c_(t)^((i))).The binary rank criterion for full diversity space-time codes in ISIchannels can thus be stated as follows.

With respect to the ISI channel binary rank criterion, it is assumedthat C is a linear L_(t)×l space-time code with underlying binary code Cof length N=L_(t)l operating in an ISI channel with L_(ISI) paths, wherel≧L_(t)L_(ISI). Also, assuming that every non-zero code word ccorresponds to a matrix c_(ISI) of full rank L_(t)L_(ISI) over thebinary field F, then, for BPSK transmission over the frequency selectivequasi-static fading channel 103, the space-time code C achieves fulltransmit diversity L_(t)L_(ISI).

While the previous result was stated for full transmit diversity codes,it readily generalizes to any order of transmit diversity less than orequal to L_(t)L_(ISI). The ISI channel binary rank criterion permits theuse of a stacking construction that establishes an algebraic frameworkfor the design of algebraic space-time codes for MIMO ISI fadingchannels. According to an embodiment of the present invention, the ISIchannel stacking construction, M₁, M₂, . . . , M_(L) _(t) are binarymatrices of dimension k×l,l≧k, and C is the L_(t)×l space-time code ofdimension k including the code word matrices

${c = \begin{bmatrix}{\underset{\_}{x}\; M_{1}} \\{\underset{\_}{x}\; M_{2}} \\\vdots \\{\underset{\_}{x}\; M_{L_{t}}}\end{bmatrix}},$

where x denotes an arbitrary k-tuple of information bits and L_(t)<l.The following is denoted

M_(n, m) = ⌊O_(L_(t) × (m − 1))M_(n)O_(L_(t) × (L_(I S I) + 1 − m))⌋,

where O_(L) _(t) _(x(m−1)) is the L_(t)×(m−1) all zero matrix. Hence, Csatisfies the ISI channel binary rank criterion, and accordingly, forBPSK transmission over the quasi-static fading channel, achieves falltransmit diversity L_(t)L_(ISI), if and only if M_(1,1), M_(2,1), . . ., M_(L) _(t) _(L) _(ISI) have the property that

∀a₁, a₂, . . . , a_(L) _(t) εF:

M=a₁M_(1,1)⊕a₂M_(2,1)⊕ . . . ⊕a_(L) _(t) _(L) _(ISI) M_(L) _(t) _(L)_(ISI) is of full rank k unless a₁= . . . a_(L) _(t) _(L) _(ISI) =0. Itis noted that

$c_{I\; S\; I} = {\begin{bmatrix}{\underset{\_}{x}\; M_{1,1}} \\{\underset{\_}{x}\; M_{1,2}} \\\vdots \\{\underset{\_}{x}\; M_{L_{t},L_{I\; S\; I}}}\end{bmatrix}.}$

The stacking construction is general and applies to block codes as wellas trellis codes. An important example of the stacking construction isgiven by the class of binary convolutional codes. This class isimportant because it allows for a reasonable complexity maximumlikelihood decoder. Let C be the binary, rate l/L_(t), convolutionalcode having transfer function matrix [6]G(D)=└g₁(D), g₂(D), . . . , g_(L) _(t) _(,1)(D), . . . , g_(L) _(t)_(,L) _(ISI) (D)┘,

then the natural space-time code C associated with C is defined toinclude the code word matrices c(D)=G^(T)(D)x(D), where the polynomialx(D) represents the input information bit stream. In other words, forthe natural space-time code, the natural transmission format is adopted,in which the output coded bits generated by g_(i)(D) are transmitted viaantenna i. It is assumed the trellis codes are terminated by tail bits[3]. Thus, if x(D) is restricted to a block of N information bits, thenC is an L_(t)×(N+v) space-time code, where v=max_(1≦i≦L) _(t) {degg_(i)(x)} is the maximal memory order of the convolutional code C. Thefollowing is denotedG _(ISI)(D)=[g_(1,1)(D), g_(2,1)(D), . . . , g_(L) _(t) _(,1)(D), . . ., g_(L) _(t) _(,L) _(ISI) (D)]

where g_(n.m)=D^((m−1))g_(n). The following characterizes the result ofthe performance of natural space-time convolutional codes in ISIchannels.

The natural space-time code C associated with the rate 1/L_(t)convolutional code C satisfies the binary rank criterion, and thusachieves full transmit diversity for BPSK transmission in an ISI channelwith L_(ISI) paths, if and only if the transfer function matrixG_(ISI)(D) of C has full rank L_(t)L_(ISI) as a matrix of coefficientsover F. This result stems from the observation that

${\sum\limits_{{1 \leq i \leq L_{t}},{1 \leq j \leq L_{I\; S\; I}}}{a_{i,j}{g_{i,j}(D)}{x(D)}}} = {{{0\mspace{14mu}{for}\mspace{14mu}{some}\mspace{14mu}{x(D)}} \neq {0\mspace{14mu}{iff}{\sum\limits_{{1 \leq i \leq L_{t}},{1 \leq j \leq L_{I\; S\; I}}}{a_{i,j}{g_{i,j}(D)}}}}} = 0.}$

This observation readily generalizes to recursive convolutional codes.

The above result extends to convolutional codes with arbitrary rates andarbitrary diversity orders. Since the coefficients of G_(ISI)(D) form abinary matrix of dimension L_(t)L_(ISI)×(v+L_(ISI)), and the column rankmust be equal to the row rank, the result provides a simple bound as tohow complex the convolutional code must be in order to satisfy the fulldiversity ISI channel binary rank criterion.

The maximum diversity order achieved by a space-time code based on anunderlying rate 1/L_(t) convolutional code C with a maximal memory orderv in a L_(ISI) paths ISI channel is v +L_(ISI). This bound shows that,for a fixed trellis complexity, increasing the number of antennas beyond

$L_{t} = \frac{v + L_{ISI}}{L_{ISI}}$will not result in an increase in the diversity advantage. This fact issupported by the results in Table 1, below, which lists the diversityadvantage for BPSK algebraic space-time codes with optimal free distancefor MIMO frequency selective fading channels:

TABLE 1 d for d for d for d for L_(t) v Connection Polynomials L_(ISI) =1 L_(ISI) = 2 L_(ISI) = 3 L_(ISI) = 4 2 2 5, 7 2 4 5 6 3 64, 74 2 4 6 74 46, 72 2 4 6 8 5 65, 57 2 4 6 8 6 554, 744 2 4 6 8 3 3 54, 64, 74 3 56 7 4 52, 66, 76 3 6 7 8 5 47, 53, 75 3 6 8 9 6 554, 624, 764 3 6 9 10 44 52, 56, 66, 76 4 6 7 8 5 53, 67, 71, 75 4 7 8 9 5 5 75, 71, 73, 65, 575 7 8 9

Because the number of paths is not known a priori at the transmitter200, it is desirable to construct space-time codes that achieve themaximum diversity order for arbitrary number of paths. This leads to thenotion of universal space-time codes that combine the maximum spatialdiversity with the ISI channel frequency diversity whenever available.Within the class of universal space-time codes with maximum diversityadvantage, it is ideal to select the code with the maximum productdistance, which measures the asymptotic coding achieved by the code [3][4].

Although BSPK modulation is discussed, it is recognized that theextension to QPSK modulation can be readily made. The ISI binary rankcriterion and stacking construction for BPSK modulation can begeneralized to obtain similar results for QPSK modulation. As aconsequence of the QPSK ISI binary rank criterion and stackingconstruction, it is observed that the binary connection polynomials ofTable 1 can be used to generate linear, Z₄-valued, rate 1/L_(t)convolutional codes whose natural space-time formatting achieves fullspatial diversity L_(t)L_(ISI) for QPSK modulation. More generally, anyset of Z₄-valued connection polynomials with modulo 2 projections (shownTable 1) may be used. In most cases under conisderation, the bestperformance was obtained from the lifted Z₄ codes constructed byreplacing the zero coefficients by twos. This lifting produces the codesin Table 2, which lists Z₄ space-time codes for QPSK modulation in MIMOfrequency selective fading channels.

TABLE 2 L₁ v Connection Polynomials 2 1 1+2D, 2+D 2 1+2D+D², 1+D+D² 31+D+2D²+D³, 1+D+D²+D³ 4 1+2D+2D²+D³+D⁴, 1+D+D²+2D³+D⁴ 51+D+2D²+D³+2D⁴+D⁵, 1+2D+D²+D³+D⁴+D⁵ 3 2 1+2D+2D², 2+D+2D², 1+D+2D² 31+D+2D²+D³, 1+D+2D²+D³, 1+D+D²+D³ 4 1+2D+D²+2D³+D⁴, 1+D+2D²+D³+D⁴,1+D+D²+D³+D⁴ 5 1+2D+2D²+D³+D⁴+D⁵, 1+2D+D²+2D³+D⁴+D⁵, 1+D+D²+D³+2D⁴+D⁵ 43 1+2D+2D²+2D³, 2+D+2D²+2D³, 2+2D+D²+2D³, 2+2D+2D²+D³ 4 1+2D+D²+2D³+D⁴,1+D+2D²+D³+D⁴, 1+D+2D²+D³+D⁴, 1+D+D²+D³+D⁴ 5 1+2D+D²+2D³+D⁴+D⁵,1+D+2D²+D³+D⁴+D⁵, 1+D+D²+2D³+2D⁴+D^(5,) 1+D+D²+D³+2D⁴+D⁵ 5 41+2D+2D²+2D³+2D⁴, 2+D+2D²+2D³+2D⁴, 2+2D+D²+2D³+2D⁴, 2+2D+2D²+D³+2D⁴,2+2D+2D²+2D³+D⁴ 5 1+D+D²+D³+2D⁴+D⁵, 1+D+D²+2D³+2D⁴+D⁵,1+D+D²+2D³+D⁴+D^(5,) 1+D+2D²+D³+2D⁴+D⁵, 1+2D+D²+D³+2D⁴+D⁵

The described single carrier time domain design approach requires theuse of a relatively more complex maximum likelihood decoder 305 toaccount for the multi-input multi-output ISI nature of the channel 103.In an exemplary embodiment, this maximum likelihood decoder 305 can berealized using a Viterbi decoder with trellis complexity proportional to2^((L) ^(ISI) ^(+v)) and 4^((L) ^(ISI) ^(+v)) for BPSK and QPSKmodulations, respectively (wherein v is the maximal memory order of theunderlying convolutional code).

If receiver complexity presents an issue, which is conceivable incertain applications, then a second design approach may be implemented.Such an approach uses space-frequency codes. In particular, to reducethe complexity of the receiver 300, an OFDM front-end 313 is utilized totransform the ISI channel into a flat, however, selective fadingchannel. The baseband signal assigned to each antenna 207 is passedthrough an inverse fast Fourier transform (IFFT) before transmission.The transmitted signal from antenna i at the nth interval is given by

${x_{n}^{i} = {\sum\limits_{k = 0}^{N - 1}{s_{k}^{i}{\exp\left( {{- j}\frac{2\pi\; k\; n}{N}} \right)}}}},$

where N is block length. A cyclic prefix of length L_(ISI)−1 is added toeliminate the ISI between consecutive OFDM symbols. At the receiver end,the signal y_(n) ^(j) received by antenna j at time t is given by

$\begin{matrix}{y_{n}^{i} = {{\sqrt{E_{s}}{\sum\limits_{l = 0}^{L_{I\; S\; I} - 1}{\sum\limits_{i = 1}^{L_{t}}{\alpha_{l}^{i\; j}x_{t - 1}^{j}}}}} + n_{t}^{j}}} \\{= {{\sqrt{E_{s}}{\sum\limits_{l = 0}^{L_{I\; S\; I} - 1}{\sum\limits_{i = 1}^{L_{t}}{\sum\limits_{k = 0}^{N - 1}{\alpha_{l}^{i\; j}s_{k}^{j}{\exp\left( {{- j}\;\frac{2\;\pi\;{k\left( {n - 1} \right)}}{N}} \right)}}}}}} + n_{t}^{j}}}\end{matrix}$

The fast Fourier transform (FFT) operator is then applied to thereceived signal to yield

$\begin{matrix}{r_{t}^{j} = {\sum\limits_{k = 0}^{N - 1}{y_{k}^{i}{\exp\left( {{- j}2\pi\; n\;\frac{t}{N}} \right)}}}} \\{= {{\sum\limits_{i = 1}^{L_{t}}{\left( {\sum\limits_{l = 0}^{L_{I\; S\; I} - 1}{\alpha_{l}^{\;{i\; j}}{\exp\left( {{- j}\;\frac{2\;\pi\; n\; t}{N}} \right)}}} \right)s_{t}^{\; i}}} + N_{t}^{j}}} \\{{= {{\sum\limits_{i = 1}^{L_{t}}{H_{t}^{({i\; j})}s_{t}^{i}}} + N_{t}^{j}}},}\end{matrix}$

where N_(t) ^(j) are independent noise samples of circularly symmetriczero-mean complex Gaussian random variable with variance N₀/2 perdimension. The complex fading coefficients of the equivalent channelmodel H_(t) ^(ij) have the following auto-correlation function:

${{R\left( {{i_{1} - i_{2}},{j_{1} - j_{2}},{t_{1} - t_{2}}} \right)} = {{E\left( {H_{t_{1}}^{({i_{1}j_{1}})}{H_{t_{2}}^{({i_{2}j_{2}})}}^{*}} \right)} = {{\delta\left( {{i_{1} - i_{2}},{j_{1} - j_{2}}} \right)}{\sum\limits_{l = 0}^{L_{I\; S\; I} - 1}{\exp\left( {{- j}\frac{2\;\pi\;{l\left( {t_{1} - t_{2}} \right)}}{N}} \right)}}}}},$

where δ(i, j) is the dirac-delta function. It is clear that the fadingcoefficients of the equivalent channel are spatially independent [6] andthat

${R\left( {0,0,\frac{kN}{L_{ISI}}} \right)} = 0$for k=1,2, . . . , L_(ISI)−1. This observation suggests that theequivalent fading channel can be approximated by the piece-wise constantblock fading channel. In this model the code word encompasses L_(ISI)fading blocks. It is assumed that the complex fading gains are constantover one fading block, but are independent from block to block. Anothertype of receiver may be utilized in the event that receiver complexitypresents a key design concern, as shown in FIG. 3B.

FIG. 3B shows a diagram of a receiver that employs space-frequencycodes, according to an embodiment of the present invention. As withreceiver 300 of the space-time code approach, receiver 311 processessignals via antennas 309 and includes a demodulator 315, a decoder 317,and a memory 319. Unlike receiver 300, receiver 311 employs an OFDMfront-end 313, and includes a fast Fourier transform (FFT) logic 321that may operate in parallel with the demodulator 315.

The design of space-frequency codes for the OFDM based design approachis described below. These space-frequency codes optimally exploit bothspatial and frequency-selective diversity available in themulti-input-multi-output (MIMO) block fading channel. As in the singlecarrier time domain design approach, attention is focused on trellisbased codes because of the availability of reasonable complexity MLdecoders. For the purpose of explanation, the discussion pertains toBPSK modulated systems; however, it is recognized by one of ordinaryskill in the art that QPSK codes can be obtained by lifting the BPSKcodes, as described previously.

The general case in which C is a binary convolutional code of ratek/L_(t)L_(ISI).is considered. The encoder 203 processes k binary inputsequences x₁(t, x₂(t), . . . , x_(k)(t) and produces L_(t)L_(ISI) codedoutput sequences y₁(t), y₂(t), . . . , y_(L) _(t) _(L) _(ISI) (t), whichare multiplexed together to form the output code word. The encoderaction is summarized by the following matrix equation

Y(D) = X(D)G(D), where  Y(D) = ⌊Y₁(D)Y₂(D)  …  Y_(L_(t)L_(I S I))(D)⌋, X(D) = [X₁(D)X₂(D)  …  X_(k)(D)], and${G(D)} = \left\lbrack \begin{matrix}{G_{1,1}(D)} & {G_{1,2}(D)} & \cdots & {G_{1,{L_{t}L_{I\; S\; I}}}(D)} \\{G_{2,1}(D)} & {G_{2,2}(D)} & \cdots & {G_{1,{L_{t}L_{I\; S\; I}}}(D)} \\\vdots & \vdots & ⋰ & \vdots \\{G_{k,1}(D)} & {G_{k,2}(D)} & \cdots & {G_{k,{L_{t}L_{I\; S\; I}}}(D)}\end{matrix} \right\rbrack$

The natural space-time formatting of C is such that the output sequencecorresponding to Y_((m−1)L) _(1+l) (D) is assigned to the l^(th)transmit antenna in the m^(th) fading block. The algebraic analysistechnique considers the rank of matrices formed by concatenating linearcombinations of the column vectors

${F_{l}(D)} = \begin{bmatrix}{G_{1,l}(D)} \\{G_{2,l}(D)} \\\vdots \\{G_{k,l}(D)}\end{bmatrix}$

G is defined to be the set of binary full rank matrices{G:G=└g_(i,j)┘_(L) _(t) _(×L) _(t) } resulting from applying any numberof simple row operations to the identity matrix I_(L) _(t) ; and ∀G₁εG,1≦i≦L_(t)1≦i≦L_(ISI),

${R_{i}^{({G_{m},m})}(D)} = {\left\lbrack {{{g_{i,1}(m)}I_{k}},{{g_{i,2}(m)}I_{k}},\ldots\mspace{11mu},{{g_{i,L_{t}}(m)}I_{k}}} \right\rbrack\begin{bmatrix}{F_{{{({m - 1})}L_{t}} + 1}(D)} \\{F_{{{({m - 1})}L_{t}} + 2}(D)} \\\vdots \\{F_{m\; L_{t}}(D)}\end{bmatrix}}$

Accordingly, the following algebraic construction for BPSKspace-frequency convolutional codes results. In a MIMO OFDM basedcommunications system with L_(t) transmit antennas 207 operating over afrequency selective block fading channel with L_(ISI) blocks, C denotesthe space-frequency code that includes the binary convolutional code C,whose k×L_(t)L_(ISI) transfer function matrix is G(D)=└F₁(D) . . . F_(L)_(t) _(L) _(ISI) (D)┘ and the spatial parser σ in which the outputY_((m−1)L) _(t) _(+l)(D)=X(D)F_((m−1)L) _(t) _(+l)(D) is assigned toantenna l in fading block m. Then, for BPSK transmission, C achieves dlevels of transmit diversity if d is the largest integer such that

${\forall{G_{1} \in G}},\ldots\mspace{11mu},{G_{L_{I\; S\; I}} \in G},{0 \leq m_{1} \leq {\min\left( {L_{t},{{L_{I\; S\; I}L_{t}} - d + 1}} \right)}},\ldots\mspace{11mu},{0 \leq m_{L_{I\; S\; I}} \leq {\min\left( {L_{t},{{L_{I\; S\; I}L_{t}} - d + 1}} \right)}},{{{and}\mspace{14mu}{\sum\limits_{i = 1}^{L_{I\; S\; I}}m_{i}}} = {{L_{I\; S\; I}L_{t}} - d + 1}},{{R_{m_{1},\ldots\mspace{11mu},{m\; L_{I\; S\; I}}}^{({G_{1},\ldots\mspace{11mu},G_{L_{I\; S\; I}}})}(D)} = \left\lbrack {{R_{0}^{({G_{1},1})}(D)},\ldots\mspace{11mu},{R_{m_{1}}^{({G_{1},1})}(D)},{R_{0}^{({G_{2},2})}(D)},\ldots\mspace{11mu},{R_{m_{2}}^{({G_{2},2})}(D)},\ldots\mspace{11mu},{R_{{m\;}_{L_{I\; S\; I}}}^{({G_{L_{I\; S\; I}},L_{I\; S\; I}})}(D)}} \right\rbrack}$has a rank k over the space of all formal series.

The above result allows for constructing convolutional space-frequencycodes that realize the optimum tradeoff between transmission rate anddiversity order for BPSK modulation with arbitrary coding rate, numberof transmit antenna, and number of fading blocks. It is readily seenthat this framework encompasses as a special case rate 1/n′convolutional codes with bit or symbol interleaving across the transmitantennas and frequency fading blocks.

Similar to the space-time coding approach, rate 1/L_(t) convolutionalcodes are considered, wherein the same transmission throughput isachieved. The output sequence from the ith arm Y_(i)(D) is assigned tothe ith antenna. The input assigned to each antenna 207 is thendistributed across the different fading blocks using a periodic bitinterleaver 209. The design of interleaver 209 depends largely onwhether the number of resolvable paths is available at the transmitter200. In the case in which this information is available at thetransmitter 200, the interleaver mapping function π is defined as

${{\pi(i)} = {\left\lbrack \frac{i}{L_{I\; S\; I}} \right\rbrack + {\frac{N}{L_{I\; S\; I}}(i)_{L_{I\; S\; I}}}}},$

where ( )_(m) refers to the modulo m operation, 0≦i≦N−1, and N is thecode word length, which is assumed to be a multiple of L_(ISI).

In the absence of the prior information on the number of resolvablepaths in the channel 103, an interleaving scheme that is capable ofexploiting all the frequency diversity, whenever available, for anarbitrary unknown number of paths is needed. In the special case inwhich the number of paths is restricted to L_(ISI)=2^(r) (for anyarbitrary integer r) and the maximum possible number of paths L_(ISI)^((max)) is known at the transmitter 200, the following construction forthe universal interleaving map is provided:

$\begin{matrix}{{{\pi(i)} = {{\sum\limits_{k = 0}^{\log_{2}{(L_{ISI}^{(\max)})}}\;{a_{k}\frac{N}{2^{k + 1}}}} + \left\lbrack \frac{i}{L_{ISI}^{(\max)}} \right\rbrack}},} \\{a_{k} = \left( \frac{{(i)L_{ISI}^{(\max)}} - {\sum\limits_{j = 0}^{k - 1}\;{a_{j}2^{j}}}}{2^{k}} \right)}\end{matrix}$

This interleaving scheme distributes the input sequence periodicallyamong the L_(ISI) fading blocks for any L_(ISI)=2^(r) andL_(ISI)≦L_(ISI) ^((max)). In practical applications, L_(ISI) ^((max))may be chosen to be larger than the maximum number of resolvable pathsexpected in this particular application, and hence, the transmitter 200does not need feedback from the receiver 300. This does not result inany loss of performance. If the number of paths is not a power of two,then the diversity advantage is lower bounded by that achieved with thenumber of paths equal to L_(ISI) ^((approx))) such that L_(ISI)^((approx))=2^(r)<L_(ISI).

Table 3 shows the diversity advantage that is achieved by the optimalfree distance codes when used as space-frequency codes in this scenario.Specifically, Table 3 lists the diversity advantage for BPSK algebraicspace-frequency codes with optimal free distance for MIMO frequencyselective fading channels.

TABLE 3 d for d for d for d for L_(t) v Connection Polynomials L_(ISI) =1 L_(ISI) = 2 L_(ISI) = 3 L_(ISI) = 4 2 2 5, 7 2 4 5 6 3 64, 74 2 4 6 74 46, 72 2 4 6 8 5 65, 57 2 4 6 8 6 554, 744 2 4 6 8 3 3 54, 64, 74 3 4— — 4 52, 66, 76 3 3 5 — 5 47, 53, 75 3 — — — 6 554, 624, 764 3 — — — 44 52, 56, 66, 76 4 — — — 5 53, 67, 71, 75 4 — — — 5 5 75, 71, 73, 65, 575 — — —

While, the codes in Table 3 may not realize the maximum possiblediversity advantage under all circumstances, these codes a compromisebetween the diversity advantage and coding gain.

As noted previously, the OFDM based approach addresses the need for alower complexity maximum likelihood receiver 300. This approachrecognizes the fact that the maximum likelihood decoder 317 complexityin the OFDM approach does not increase exponentially with the number ofresolvable paths, contrary to the space-time coding approach. It shouldbe noted that this does not mean, however, that complexity of thedecoder 317 does not depend on the number of paths. As shown in Table 3,as the number of paths increases, the codes with larger constraintlengths are needed to efficiently exploit the diversity available in thechannel 103. Unlike the space-time coding approach, it is possible totrade diversity advantage for a reduction in complexity by choosing acode with a small constraint length. This trade-off is not possible inthe space-time coding approach because, irrespective of the constraintlength of the code, the complexity of the (ML) decoder 305 growsexponentially with the number of resolvable paths. The OFDM basedapproach, however, provides a relatively lower diversity advantage overthe space-time coding approach.

The maximum transmit diversity advantage achieved in a BPSK OFDM MIMOwireless system with L_(t) transmit antennas 207 and L_(ISI) resolvablepaths/antenna supporting a throughput of 1 bps/Hz is L_(ISI)(L_(t)−1)+1.It is clear that the maximum diversity advantage under this approach islower as compared to the space-time coding approach (i.e, L_(t)L_(ISI)).The results in Tables 1 and 3 compare the diversity advantage achievedby space-time codes and space-frequency codes for different values ofL_(t) and L_(ISI). As will be evident from the discussion below, thisloss in diversity advantage may not always lead to a performance loss inthe frame error rate range of interest.

FIG. 4 shows a diagram of a communications system that supportsmultichannel multipoint distribution services using multi-inputmulti-output transceivers, according to the embodiments of the presentinvention. The present invention provides a physical layer (PHY) forbroadband fixed wireless access, as supported by the system 400. A basestation 401 communicates with multiple terminals 403, 405 over acommunications channel 407, which in this example is characterized bymultipath fading. The system 400 exploits the additional temporal (orfrequency) diversity available in the channel 407, thereby improvingdata rates. Each of the terminals 403, 405 as well as the base station401 is equipped with an encoder 203 (as shown in FIG. 2) that generatesspace-time (ST) or space-frequency codes. ST coding is well-suited toadapting to subscriber density variation. In an exemplary embodiment,the error rate, after application of the appropriate error correctionmechanism (e.g., ARQ), that is delivered by the PHY layer to a MediumAccess Control (MAC) layer satisfies IEEE (Institute of Electrical andElectronics Engineers) 802 functional requirements, e.g., bit error rate(BER) is 10⁻⁹. The base station 401 transmits parallel data streams tothe terminals 403, 405 using OFDM with QPSK (TDD or FDD), along withARQ. It is noted that Quadrature Amplitude Modulation (QAM) may also beused.

OFDM handles delay spread with low complexity at the receiver. OFDMtransforms frequency-selective multi-path channel into block fadingchannel, a highly efficient way to deal with multi-paths. For apractical delay spread, the complexity can be significantly lower thanthat of a high-speed, single carrier system with an equalizer. The MIMOtransceivers of the base station 401 use space-time codes in LOSenvironment to increase capacity, which is dependent on the number ofantennas. The system 400 can also be used in a near or non-LOS by takingadvantage of the multipath characteristics of the channel 407 (e.g.,assuming the frequency is between about 2-5 GHz). Conventionalapproaches for line-of-sight (LOS) coverage provides only limitedcapacity, limited availability, and is potentially difficult to installthe cell site (i.e., base station). Highly sectored solutions areimpractical due to propagation environment and non-uniform subscriberdistribution; high-order modulation are sensitive to multipath and otherimpairments.

In an exemplary embodiment, the return channel to the base station 401is a low power, narrower band. It is noted that an asymmetric number ofantennas from the base station 401 to terminal 403, 405 can be used, inpart, to reduce cost of the customer premises equipment. Accordingly,the MIMO architecture may provide a single transmit antenna and tworeceive antennas at the terminals 403, 405; alternatively, theseterminals 403, 405 may have a configuration in two transmit antennas andtwo receive antennas are used. At the base station 401, four transmitantennas and four receive antennas may be implemented, as shown. Toincrease capacity, different polarizations (i.e., left and right) aresupported by the system 400 in the frequency reuse pattern.

It is recognized that the terminals 403, 405 and the base station 401maybe configured to operate using space-frequency codes. As mentionedpreviously, the choice of space-time codes versus space-frequency codesdepends largely on the trade-off between receiver complexity and thedesired diversity advantage.

Unlike the conventional approach to providing fixed wireless access,which operates only with LOS coverage with limited capacity andavailability, the system 400 supports high capacity and availability,thereby providing a via alternative to high speed terrestrial accesstechnologies, such as Digital Subscriber Line (DSL). The system 400advantageously permits more flexibility in the trade-off between rateand diversity advantage.

FIG. 5 shows a graph of the performance of a four transmit/four receiveantenna system employed in the system of FIG. 4, according toembodiments of the present invention. In a four transmit antenna/fourreceive antenna configuration, as used in the base station 401 (FIG. 4),for a given Frame Error Rate (FER), the conventional system requires ahigher Carrier to Noise Ratio (CNR)—i.e., over 3 dBs greater. The framesize, in an exemplary embodiment, is 256 information bits; a desirableFER is 1%, and a bandwidth of 6 Mhz. For example, a FER of 0.01 requiresabout 3.5 dBs using the coding and modulation scheme of the system 400,while the conventional system needs over 6.5 dBs to attain an equivalentFER. As a broadband system, the system 400 implies large delay spreadrelative to transmission symbol interval. For example, at 6 MHztransmission rate, a delay spread of 10 msec extends over 60 symbols.The performance parameters of FIG. 5 are achieved using Binary PhaseShift Keying (BPSK). FIG. 6 shows the FER versus CNR for an antennaconfiguration that may be deployed in the terminals 403, 405 of thesystem 400.

FIG. 6 shows a graph of the performance of a two transmit/one receiveantenna system, employed in the system of FIG. 4, according toembodiments of the present invention. The results of FIG. 6 assumes aQPSK modulation scheme. As shown, CNR improvements become greater as theFER gets reduced. At an FER of 0.01, the system 400 shows an improvementof 1 dB.

Accordingly, the system 400 may support fixed wireless radio channelsfor various data applications (e.g., Internet access). Theseapplications are particularly suitable for small office/home office(SOHO) environments.

FIG. 7 illustrates a computer system 700 upon which an embodimentaccording to the present invention can be implemented. The computersystem 700 includes a bus 701 or other communication mechanism forcommunicating information, and a processor 703 coupled to the bus 701for processing information. The computer system 700 also includes mainmemory 705, such as a random access memory (RAM) or other dynamicstorage device, coupled to the bus 701 for storing information andinstructions to be executed by the processor 703. Main memory 705 canalso be used for storing temporary variables or other intermediateinformation during execution of instructions to be executed by theprocessor 703. The computer system 700 further includes a read onlymemory (ROM) 707 or other static storage device coupled to the bus 701for storing static information and instructions for the processor 703. Astorage device 709, such as a magnetic disk or optical disk, isadditionally coupled to the bus 701 for storing information andinstructions.

The computer system 700 may be coupled via the bus 701 to a display 711,such as a cathode ray tube (CRT), liquid crystal display, active matrixdisplay, or plasma display, for displaying information to a computeruser. An input device 713, such as a keyboard including alphanumeric andother keys, is coupled to the bus 701 for communicating information andcommand selections to the processor 703. Another type of user inputdevice is cursor control 715, such as a mouse, a trackball, or cursordirection keys for communicating direction information and commandselections to the processor 703 and for controlling cursor movement onthe display 711.

According to one embodiment of the invention, the space-time encoding isprovided by the computer system 700 in response to the processor 703executing an arrangement of instructions contained in main memory 705.Such instructions can be read into main memory 705 from anothercomputer-readable medium, such as the storage device 709. Execution ofthe arrangement of instructions contained in main memory 705 causes theprocessor 703 to perform the process steps described herein. One or moreprocessors in a multi-processing arrangement may also be employed toexecute the instructions contained in main memory 705. In alternativeembodiments, hard-wired circuitry may be used in place of or incombination with software instructions to implement the embodiment ofthe present invention. Thus, embodiments of the present invention arenot limited to any specific combination of hardware circuitry andsoftware.

The computer system 700 also includes a communication interface 717coupled to bus 701. The communication interface 717 provides a two-waydata communication coupling to a network link 719 connected to a localnetwork 721. For example, the communication interface 717 may be adigital subscriber line (DSL) card or modem, an integrated servicesdigital network (ISDN) card, a cable modem, or a telephone modem toprovide a data communication connection to a corresponding type oftelephone line. As another example, communication interface 717 may be alocal area network (LAN) card (e.g. for Ethernet™ or an AsynchronousTransfer Model (ATM) network) to provide a data communication connectionto a compatible LAN. Wireless links can also be implemented. In any suchimplementation, communication interface 717 sends and receiveselectrical, electromagnetic, or optical signals that carry digital datastreams representing various types of information. Further, thecommunication interface 717 can include peripheral interface devices,such as a Universal Serial Bus (USB) interface, a PCMCIA (PersonalComputer Memory Card International Association) interface, etc. Althoughonly a single communication interface 717 is shown, it is recognizedthat multiple communication interfaces may be employed to communicatewith different networks and devices.

The network link 719 typically provides data communication through oneor more networks to other data devices. For example, the network link719 may provide a connection through local network 721 to a hostcomputer 723, which has connectivity to a network 725 (e.g. a wide areanetwork (WAN) or the global packet data communication network nowcommonly referred to as the “Internet”) or to data equipment operated byservice provider. The local network 721 and network 725 both useelectrical, electromagnetic, or optical signals to convey informationand instructions. The signals through the various networks and thesignals on network link 719 and through communication interface 717,which communicate digital data with computer system 700, are exemplaryforms of carrier waves bearing the information and instructions.

The computer system 700 can send messages and receive data, includingprogram code, through the network(s), network link 719, andcommunication interface 717. In the Internet example, a server (notshown) might transmit requested code belonging to an application programfor implementing an embodiment of the present invention through thenetwork 725, local network 721 and communication interface 717. Theprocessor 703 may execute the transmitted code while being receivedand/or store the code in storage device 79, or other non-volatilestorage for later execution. In this manner, computer system 700 mayobtain application code in the form of a carrier wave.

The term “computer-readable medium” as used herein refers to any mediumthat participates in providing instructions to the processor 703 forexecution. Such a medium may take many forms, including but not limitedto non-volatile media, volatile media. Non-volatile media include, forexample, optical or magnetic disks, such as storage device 709. Volatilemedia include dynamic memory, such as main memory 705. Common forms ofcomputer-readable media include, for example, a floppy disk, a flexibledisk, hard disk, magnetic tape, any other magnetic medium, a CD-ROM,CDRW, DVD, any other optical medium, punch cards, paper tape, opticalmark sheets, any other physical medium with patterns of holes or otheroptically recognizable indicia, a RAM, a PROM, and EPROM, a FLASH-EPROM,any other memory chip or cartridge, or any other medium from which acomputer can read.

Various forms of computer-readable media may be involved in providinginstructions to a processor for execution. For example, the instructionsfor carrying out at least part of the present invention may initially beborne on a magnetic disk of a remote computer. In such a scenario, theremote computer loads the instructions into main memory and sends theinstructions over a telephone line using a modem. A modem of a localcomputer system receives the data on the telephone line and uses aninfrared transmitter to convert the data to an infrared signal andtransmit the infrared signal to a portable computing device, such as apersonal digital assistance (PDA) and a laptop. An infrared detector onthe portable computing device receives the information and instructionsborne by the infrared signal and places the data on a bus. The busconveys the data to main memory, from which a processor retrieves andexecutes the instructions. The instructions received by main memory mayoptionally be stored on storage device either before or after executionby processor.

Accordingly, the present invention provides a MMDS system that employsspace-time codes to transmit and receives signals under LOS and non-LOSoperating conditions via MIMO transceivers. The system utilizesorthogonal frequency division multiplexing (OFDM) with a quadraturephase shift keying (QPSK) sub-carrier modulation, according to oneembodiment of the present invention. The space-time coding, which may beadaptive, effectively exploits the use of the MIMO transceivers.Further, the MMDS system provides an automatic repeat requests (ARQ)error control mechanism.

While the present invention has been described in connection with anumber of embodiments and implementations, the present invention is notso limited but covers various obvious modifications and equivalentarrangements, which fall within the purview of the appended claims.

REFERENCES

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[3] V. Tarokh, N. Seshadri, and A. R. Calderbank. Space-Time Codes forHigh Data Rate Wireless Communication: Performance Criterion and CodeConstruction. IEEE Trans. Info. Theory, IT-44:774-765, March 1998.

[4] J.-C. Guey, M. R. Bell M. P. Fitz, and W.-Y. Kuo. Signal Design forTransmitter Diversity, Wireless Communications systems over RayleighFading Channels. IEEE Vehicular Technology Conference, pages 136-140,Atlanta, 1996.

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[6] S. Lin and Jr. D. J. Costello. Error Control Coding: Fundamentalsand Applications. Prentice-Hall, N.J., 1983.

1. A method for communicating over a radio communications system, themethod comprising: encoding a signal using space-frequency coding byutilizing a transfer function matrix that is an element of a set ofbinary full-rank matrices whose structure is defined by number of pathsin an intersymbol Interference (ISI) channel and number of transmitantennas; modulating the encoded signal according to a predeterminedmodulation scheme and an orthogonal frequency division multiplexingscheme; and transmitting the modulated signal to a terminal using aplurality of multiple-input multiple-output (MIMO) transceivers.
 2. Themethod according to claim 1, wherein the predetermined modulation schemeincludes at least of one Quadrature Amplitude Modulation (QAM) andQuadrature Phase Shift Keying (QPSK).
 3. The method according to claim1, wherein the encoding step is adaptive based on quantity of theplurality of MIMO transceivers.
 4. The method according to claim 1,wherein the transmitting step is performed according to an error controlscheme that includes automatic repeat request (ARQ).
 5. The methodaccording to claim 1, wherein the plurality of transceivers in thetransmitting step are coupled to a plurality of antennas that are as atleast one of a set of two transmit antennas and two receive antennas, aset of four transmit antennas and four receive antennas, and a set offour transmit antennas and two receive antennas.
 6. The method accordingto claim 1, wherein the encoding step is based on a rate ½ trellis-basedcode.
 7. A device for communicating over a radio communications system,the device comprising: an encoder configured to encode a signal usingspace-frequency coding by utilizing a transfer function matrix that isan element of a set of binary full-rank matrices whose structure isdefined by number of paths in an intersymbol interference (ISI) channeland number of transmit antennas; a modulator configured to modulate theencoded signal according to a predetermined modulation scheme and anorthogonal frequency division multiplexing scheme; and a plurality ofmultiple-input multiple-output (MIMO) transceivers coupled to themodulator, each of the transceivers being configured to transmit themodulated signal to a terminal.
 8. The device according to claim 7,wherein the predetermined modulation scheme includes at least of oneQuadrature Amplitude Modulation (QAM) and Quadrature Phase Shift Keying(QPSK).
 9. The device according to claim 7, wherein the encoderadaptively encodes based on quantity of the plurality of MIMOtransceivers.
 10. The device according to claim 7, wherein the signal istransmitted according to an error control scheme that includes automaticrepeat request (ARQ).
 11. The device according to claim 7, furthercomprising: a plurality of antennas coupled to the plurality oftransceivers the plurality of antennas comprising at least one of a setof two transmit antennas and two receive antennas, a set of fourtransmit antennas and four receive antennas, and a set of four transmitantennas and two receive antennas.
 12. The device according to claim 7,wherein the encoder encodes based on a rate ½ trellis-based code.
 13. Acommunications system comprising: a base station transmitting a signalthat is modulated according to a predetermined modulation scheme and anorthogonal frequency division multiplexing scheme, wherein the signal isencoded using space-frequency coding associated with a transfer functionmatrix that is an element of a set of binary full-rank matrices whosestructure is defined by number of paths in an intersymbol interference(ISI) channel and number of transmit antennas, the base stationincluding a plurality of multiple-input multiple-output (MIMO)transceivers; and a terminal receiving the modulated signal.
 14. Thesystem according to claim 13, wherein the predetermined modulationscheme includes at least of one Quadrature Amplitude Modulation (QAM)and Quadrature Phase Shift Keying (QPSK).
 15. The system according toclaim 13, wherein the base station adaptively encodes based an quantityof the plurality of MIMO transceivers.
 16. The system according to claim13, wherein the signal is transmitted according to an error controlscheme that includes automatic repeat request (ARQ).
 17. The systemaccording to claim 13, wherein the base station includes at least one ofa set of two transmit antennas and two receive antennas, a set of fourtransmit antennas and four receive antennas, and a set of four transmitantennas and two receive antennas, the terminal including two transmitantennas and two receive antennas.
 18. The system according to claim 13,wherein the base station encodes based on a rate ½ trellis-based code.19. An apparatus for communicating over a radio communications system,the apparatus comprising: means for encoding a signal usingspace-frequency coding by utilizing a transfer function matrix that isan element of a set of binary full-rank matrices whose structure isdefined by number of paths in an intersymbol interference (ISI) channeland number of transmit antennas; means for modulating the encoded signalaccording to a predetermined modulation scheme and an orthogonalfrequency division multiplexing scheme; and means for transmitting themodulated signal to a terminal using a plurality of multiple-inputmultiple-output (MIMO) transceivers.
 20. The apparatus according toclaim 19, wherein the predetermined modulation scheme includes at leastof one Quadrature Amplitude Modulation (QAM) and Quadrature Phase ShiftKeying (QPSK).
 21. The apparatus according to claim 19, wherein theencoding means adaptively encodes based on quantity of the plurality ofMIMO transceivers.
 22. The apparatus according to claim 19, wherein thesignal is transmitted according to an error control scheme that includesautomatic repeat request (ARQ).
 23. The apparatus according to claim 19,wherein the plurality of transceivers are coupled to a plurality ofantennas the plurality of antennas comprising at least one of a set oftwo transmit antennas and two receive antennas, a set of four transmitantennas and four receive antennas, and a set of four transmit antennasand two receive antennas.
 24. The apparatus according to claim 19,wherein the encoding means encodes based on a rate ½ trellis-based code.25. A computer-readable storage medium for storing one or more sequencesof one or more instructions for communicating over a radiocommunications system, the one or more sequences of one or moreinstructions including instructions which, when executed by one or moreprocessors, cause the one or more processors to perform the step of:encoding a signal using space-frequency coding by utilizing a transferfunction matrix that is an element of a set of binary full-rank matriceswhose structure is defined by number of paths in an intersymbolinterference (ISI) channel and number of transmit antennas, the encodedsignal being modulated according to a predetermined modulation schemeand an orthogonal frequency division multiplexing scheme, wherein themodulated signal is transmitted a terminal using a plurality ofmultiple-input multiple-output (MIMO) transceivers.
 26. Thecomputer-readable storage medium according to claim 25, wherein thepredetermined modulation scheme includes at least of one QuadratureAmplitude Modulation (QAM) and Quadrature Phase Shift Keying (QPSK). 27.The computer-readable storage medium according to claim 25, wherein theencoding step is adaptive based on quantity of the plurality of MIMOtransceivers.
 28. The computer-readable storage medium according toclaim 25, wherein the signal is transmitted according to an errorcontrol scheme that includes automatic repeat request (ARQ).
 29. Thecomputer-readable storage medium according to claim 25, wherein theplurality of transceivers are coupled to a plurality of antennas thatare as at least one of a set of two transmit antennas and two receiveantennas, a set of four transmit antennas and four receive antennas, anda set of four transmit antennas and two receive antennas.
 30. Thecomputer-readable storage medium according to claim 25, wherein theencoding step is based on a rate ½ trellis-based code.
 31. A method ofcommunicating in a multichannel multipoint distribution service (MMDS)system that includes a plurality of transmit antennas, the methodcomprising: receiving an encoded signal representing space-time codes;and transforming an intersymbol interference (ISI) channel that ischaracteristic of a channel supporting transmission of the encodedsignal into a flat black fading channel, wherein the space-time codesare generated based on a set of binary full-rank matrices whosestructure is defined by number of paths in the ISI channel and number oftransmit antennas, the space-time codes exploiting diversity associatedwith the flat block fading channel.